Datasheet

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Calculating the Current Sense Filter Network
L
To ISNS pin
UDG−04150
To VO pin
V
O
C
O
R
LDC
C
FLT
V
IN
R
FLT
100 Ω
R
FLT
+
L
R
LDC
C
FLT
* 100 (W)
(6)
Compensation for Inductor Resistance Change Over Temperature
TPS40100
SLUS601–MAY 2005
APPLICATION INFORMATION (continued)
The TPS 40100 gets current feedback information by sensing the voltage across the inductor resistance, R
LDC
.In
order to do this, a filter must be constructed that allows the sensed voltage to be representative of the actual
current in the inductor. This filter is a series R-C network connected across the inductor as shown in Figure 3.
Figure 3. Current Sensing Filter Circuit
If the R
FLT
-C
FLT
time constant is matched to the L/R
LDC
time constant, the voltage across C
FLT
is equal to the
voltage across R
LDC
. It is recommended to keep R
FLT
10 kΩ or less. C
FLT
can be arbitrarily chosen to meet this
condition (100 nF is suggested). R
FLT
can then be calculated.
where
• R
FLT
is the current sense filter resistance (Ω)
• C
FLT
is the current sense filter capacitance (F)
• L is the output inductance (H)
• R
LDC
is the DC resistance of the output inductor (Ω)
When laying out the board, better performance can be accomplished by locating C
FLT
as close as possible to the
VO and ISNS pins. The closer the two resistors can be brought to the device the better as this reduces the
length of high impedance runs that are susceptible to noise pickup. The 100-Ω resistor from V
OUT
to the VO pin
of the device is to limit current in the event that the output voltage dips below ground when a short is applied to
the output of the converter.
The resistance in the inductor that is sensed is the resistance of the copper winding. This value changes over
temperature and has approximately a 4000 ppm/°C temperature coefficient. The gain of current sense amplifier
in the TPS40100 has a built in temperature coefficient of approximately -2000 ppm/°C. If the circuit is physically
arranged so that there is good thermal coupling between the inductor and the device, the thermal shifts tend to
offset. If the thermal coupling is perfect, the net temperature coefficient is 2000 ppm/°C. If the coupling is not
perfect, the net temperature coefficient lies between 2000 ppm/°C and 4000 ppm/°C. For most applications this is
sufficient. If desired, the temperature drifts can be compensated for. The following compensation scheme
assumes that the temperature rise at the device is directly proportional to the temperature rise at the inductor. If
this is not the case, compensation accuracy suffers. Also, there is generally a time lag in the temperature rise at
the device vs. at the inductor that could introduce transient errors beyond those predicted by the compensation.
Also, the 100-Ω resistor in Figure 3 is not shown. However, it is required if the output voltage can dip below
ground during fault conditions. The calculations are not afffected, other than increasing the effective value of R
F1
by 100-Ω.
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