Datasheet

OPA843
14
SBOS268C
www.ti.com
the phase margin approaches 90°, as it does in high-gain
configurations. At low signal gains, most amplifiers will ex-
hibit a more complex response with lower phase margin. The
OPA843 is optimized to give a maximally flat 2nd-order
Butterworth response in a gain of 5. In this configuration, the
OPA843 has approximately 60° of phase margin and will
show a typical 3dB bandwidth of 260MHz. When the phase
margin is 60°, the closed-loop bandwidth is approximately
2
greater than the value predicted by dividing GBP by the noise
gain. Increasing the gain will cause the phase margin to
approach 90° and the bandwidth to more closely approach
the predicted value of (GBP/NG). At a gain of +20, the
40MHz bandwidth shown in the Electrical Characteristics
agrees with that predicted using the simple formula and the
typical GBP of 800MHz.
LOW GAIN OPERATION
Decreasing the operating gain for the OPA843 from the
nominal design point of +5 will decrease the phase margin.
This will increase the Q for the closed-loop poles, peak up
the frequency response, and extend the bandwidth. A peaked
frequency response will show overshoot and ringing in the
pulse response as well as a higher integrated output noise.
Operating at a noise gain less than +3 runs the risk of
sustained oscillation (loop instability). However, operation at
low gains would be desirable to take advantage of the much
higher slew rate and lower input noise voltage available in
the OPA843, as compared to the performance offered by
unity-gain stable op amps. Numerous external compensation
techniques have been suggested for operating a high-gain
op amp at low gains. Most of these give zero/pole pairs in the
closed-loop response that cause long term settling tails in the
pulse response and/or phase nonlinearity in the frequency
response. Figure 10 shows an external compensation method
for a noninverting configuration that does not suffer from
these drawbacks.
tune the flatness by adjusting R
I
. The Typical Characteristics
show a signal gain of +4 with the noise gain adjusted for
flatness using different values for R
1
.
Where low gain is desired, and inverting operation is accept-
able, a new external compensation technique may be used to
retain the full slew rate and noise benefits of the OPA843 while
maintaining the increased loop gain and the associated im-
provement in distortion offered by the decompensated archi-
tecture. This technique shapes the noise gain for good stability
while giving an easily controlled 2nd-order low-pass frequency
response. Figure 11 shows this circuit. Considering only the
noise gain for the circuit of Figure 11, the low-frequency noise
gain (NG
1
) will be set by the resistor ratios while the high-
frequency noise gain (NG
2
) will be set by the capacitor ratios.
The capacitor values set both the transition frequencies and
the high-frequency noise gain. If this noise gain, determined by
NG
2
= 1 + C
S
/C
F
, is set to a value greater than the recom-
mended minimum stable gain for the op amp and the noise
gain pole (set by 1/R
F
C
F
) is placed correctly, a very well
controlled 2nd-order low-pass frequency response will result.
The R
1
resistor across the two inputs will increase the noise
gain (i.e., decrease the loop gain) without changing the
signal gain. This approach will retain the full slew rate to the
output but will give up some of the low-noise benefit of the
OPA843. Assuming a low source impedance, set R
1
so that
1 + R
F
/(R
G
|| R
I
) is +3. This approach may also be used to
To choose the values for both C
S
and C
F
, two parameters
and only three equations need to be solved. The first param-
eter is the target high-frequency noise gain, NG
2
, which
should be greater than the minimum stable gain for the
OPA843. Here, a target NG
2
of 7.5 will be used. The second
parameter is the desired low-frequency signal gain, which
also sets the low-frequency noise gain, NG
1
. To simplify this
discussion, we will target a maximally flat 2nd-order low-pass
Butterworth frequency response (Q = 0.707). The signal gain
of 2 shown in Figure 11 will set the low-frequency noise gain
to NG
1
= 1 + R
F
/R
G
(= 3 in this example). Then, using only
these two gains and the GBP for the OPA843 (800MHz), the
key frequency in the compensation is determined by:
Z
GBP
NG
NG
NG
NG
NG
0
2
1
2
1
2
1
112=−
(11)
Physically, this Z
0
(13.6MHz for the values shown in Figure 11)
is set by 1/(2π R
F
(C
F
+ C
S
)) and is the frequency at which
the rising portion of the noise gain would intersect unity gain
if projected back to 0dB gain. The actual zero in the noise gain
OPA843
+5V
+5V
R
S
50
V
O
V
I
R
G
402
R
1
133
R
T
50
R
F
402
50Load
50Source
OPA843
+5V
V
O
5V
280
V
1
402
R
F
806
C
S
12.6pF
0.1µF
C
F
1.9pF
Power-supply
decoupling not shown.
R
S
= 0
FIGURE 10. Noninverting Low Gain Circuit.
FIGURE 10. Noninverting Low Gain Circuit.