Datasheet

"#$
SBOS263F − AUGUST 2004 − REVISED AUGUST 2008
www.ti.com
23
MACROMODEL AND APPLICATIONS
SUPPORT
Computer simulation of circuit performance using SPICE
is often a quick way to analyze the performance of the
OPA830 and its circuit designs. This is particularly true for
video and RF amplifier circuits where parasitic
capacitance and inductance can play a major role on
circuit performance. A SPICE model for the OPA830 is
available through the TI web page (www.ti.com). The
applications department is also available for design
assistance. These models predict typical small signal AC,
transient steps, DC performance, and noise under a wide
variety of operating conditions. The models include the
noise terms found in the electrical specifications of the
data sheet. These models do not attempt to distinguish
between the package types in their small-signal AC
performance.
OPERATING SUGGESTIONS
OPTIMIZING RESISTOR VALUES
Since the OPA830 is a unity-gain stable, voltage-feedback
op amp, a wide range of resistor values may be used for
the feedback and gain setting resistors. The primary limits
on these values are set by dynamic range (noise and
distortion) and parasitic capacitance considerations. For a
noninverting unity-gain follower application, the feedback
connection should be made with a direct short.
Below 200Ω, the feedback network will present additional
output loading which can degrade the harmonic distortion
performance of the OPA830. Above 1kΩ, the typical
parasitic capacitance (approximately 0.2pF) across the
feedback resistor may cause unintentional band limiting in
the amplifier response.
A good rule of thumb is to target the parallel combination
of R
F
and R
G
(see Figure 3) to be less than about 400Ω.
The combined impedance R
F
|| R
G
interacts with the
inverting input capacitance, placing an additional pole in
the feedback network, and thus a zero in the forward
response. Assuming a 2pF total parasitic on the inverting
node, holding R
F
|| R
G
< 400Ω will keep this pole above
200MHz. By itself, this constraint implies that the feedback
resistor R
F
can increase to several kΩ at high gains. This
is acceptable as long as the pole formed by R
F
and any
parasitic capacitance appearing in parallel is kept out of
the frequency range of interest.
In the inverting configuration, an additional design
consideration must be noted. R
G
becomes the input
resistor and therefore the load impedance to the driving
source. If impedance matching is desired, R
G
may be set
equal to the required termination value. However, at low
inverting gains, the resultant feedback resistor value can
present a significant load to the amplifier output. For
example, an inverting gain of 2 with a 50Ω input matching
resistor (= R
G
) would require a 100Ω feedback resistor,
which would contribute to output loading in parallel with the
external load. In such a case, it would be preferable to
increase both the R
F
and R
G
values, and then achieve the
input matching impedance with a third resistor to ground
(see Figure 9). The total input impedance becomes the
parallel combination of R
G
and the additional shunt
resistor.
BANDWIDTH vs GAIN:
NONINVERTING OPERATION
Voltage-feedback op amps exhibit decreasing closed-loop
bandwidth as the signal gain is increased. In theory, this
relationship is described by the Gain Bandwidth Product
(GBP) shown in the specifications. Ideally, dividing GBP
by the noninverting signal gain (also called the Noise Gain,
or NG) will predict the closed-loop bandwidth. In practice,
this only holds true when the phase margin approaches
90°, as it does in high-gain configurations. At low gains
(increased feedback factors), most amplifiers will exhibit a
more complex response with lower phase margin. The
OPA830 is compensated to give a slightly peaked
response in a noninverting gain of 2 (see Figure 3). This
results in a typical gain of +2 bandwidth of 110MHz, far
exceeding that predicted by dividing the 110MHz GBP by
2. Increasing the gain will cause the phase margin to
approach 90° and the bandwidth to more closely approach
the predicted value of (GBP/NG). At a gain of +10, the
11MHz bandwidth shown in the Electrical Characteristics
agrees with that predicted using the simple formula and
the typical GBP of 110MHz.
Frequency response in a gain of +2 may be modified to
achieve exceptional flatness simply by increasing the
noise gain to 3. One way to do this, without affecting the +2
signal gain, is to add an 2.55kΩ resistor across the two
inputs, as shown in Figure 7. A similar technique may be
used to reduce peaking in unity-gain (voltage follower)
applications. For example, by using a 750Ω feedback
resistor along with a 750Ω resistor across the two op amp
inputs, the voltage follower response will be similar to the
gain of +2 response of Figure 2. Further reducing the value
of the resistor across the op amp inputs will further dampen
the frequency response due to increased noise gain. The
OPA830 exhibits minimal bandwidth reduction going to
single-supply (+5V) operation as compared with ±5V. This
minimal reduction is because the internal bias control
circuitry retains nearly constant quiescent current as the
total supply voltage between the supply pins is changed.
INVERTING AMPLIFIER OPERATION
All of the familiar op amp application circuits are available
with the OPA830 to the designer. See Figure 9 for a typical
inverting configuration where the I/O impedances and
signal gain from Figure 1 are retained in an inverting circuit
configuration. Inverting operation is one of the more
common requirements and offers several performance
benefits. It also allows the input to be biased at V
S
/2
without any headroom issues. The output voltage can be
independently moved to be within the output voltage range
with coupling capacitors, or bias adjustment resistors.