Datasheet

ManualsBrandsTEXAS INSTRUMENTS ManualsLinear ICsLinear IC - Op-amp Voltage feedback SOT 23 5

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SBOS303C − JUNE 2004 − REVISED AUGUST 2008

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−

Frequency (Hz)

Gain (dB)

100k 1M 10M 100M

Figure 11. High-Q 1MHz Bandpass Filter

Frequency Response

DESIGN-IN TOOLS

DEMONSTRATION FIXTURES

Two printed circuit boards (PCBs) are available to assist in the

initial evaluation of circuit performance using the OPA820 in its

two package options. Both of these are offered free of charge

as unpopulated PCBs, delivered with user’s guide. The

summary information for these fixtures is shown in the table

below.

PRODUCT PACKAGE

ORDERING

NUMBER

LITERATURE

NUMBER

OPA820ID SO-8 DEM-OPA-SO-1A SBOU009

OPA820IDBV SOT23-5 DEM-OPA-SOT-1A SBOU010

The demonstration fixtures can be requested at the Texas

Instruments web site (www.ti.com) through the OPA820

product folder.

MACROMODELS AND APPLICATIONS

SUPPORT

Computer simulation of circuit performance using SPICE is

often a quick way to analyze the performance of the OPA820

and its circuit designs. This is particularly true for video and R

amplifier circuits where parasitic capacitance and inductance

can play a major role on circuit performance. A SPICE model

for the OPA820 is available through the TI web page

(www.ti.com). The applications department is also available

for design assistance. These models predict typical

small-signal AC, transient steps, DC performance, and noise

under a wide variety of operating conditions. The models

include the noise terms found in the electrical specifications of

the data sheet. These models do not attempt to distinguish

between the package types in their small-signal AC

performance.

OPERATING SUGGESTIONS

OPTIMIZING RESISTOR VALUES

Since the OPA820 is a unity-gain stable, voltage-feedback op

amp, a wide range of resistor values may be used for the

feedback and gain-setting resistors. The primary limits on

these values are set by dynamic range (noise and distortion)

and parasitic capacitance considerations. Usually, the feed-

back resistor value should be between 200Ω and 1kΩ. Below

200Ω, the feedback network will present additional output

loading which can degrade the harmonic distortion perfor-

mance of the OPA820. Above 1kΩ, the typical parasitic

capacitance (approximately 0.2pF) across the feedback

resistor may cause unintentional band limiting in the amplifier

response. A direct short is suggested as a feedback for

= +1V/V.

A good rule of thumb is to target the parallel combination of R

and R

(see Figure 1) to be less than about 200Ω. The

combined impedance R

|| R

interacts with the inverting input

capacitance, placing an additional pole in the feedback

network, and thus a zero in the forward response. Assuming

a 2pF total parasitic on the inverting node, holding R

|| R

200Ω will keep this pole above 400MHz. By itself, this

constraint implies that the feedback resistor R

can increase

to several kΩ at high gains. This is acceptable as long as the

pole formed by R

and any parasitic capacitance appearing in

parallel is kept out of the frequency range of interest.

In the inverting configuration, an additional design

consideration must be noted. R

becomes the input resistor

and therefore the load impedance to the driving source. If

impedance matching is desired, R

may be set equal to the

required termination value. However, at low inverting gains,

the resulting feedback resistor value can present a significant

load to the amplifier output. For example, an inverting gain of

2 with a 50Ω input matching resistor (= R

) would require a

100Ω feedback resistor, which would contribute to output

loading in parallel with the external load. In such a case, it

would be preferable to increase both the R

and R

values,

and then achieve the input matching impedance with a third

resistor to ground (see Figure 2). The total input impedance

becomes the parallel combination of R

and the additional

shunt resistor.

BANDWIDTH vs GAIN

Voltage-feedback op amps exhibit decreasing closed-loop

bandwidth as the signal gain is increased. In theory, this

relationship is described by the GBP shown in the

specifications. Ideally, dividing GBP by the noninverting signal

gain (also called the noise gain, or NG) will predict the

closed-loop bandwidth. In practice, this only holds true when

the phase margin approaches 90°, as it does in high-gain

configurations. At low signal gains, most amplifiers will exhibit

a more complex response with lower phase margin. The

OPA820 is optimized to give a maximally-flat, 2nd-order

Butterworth response in a gain of 2. In this configuration, the

OPA820 has approximately 64° of phase margin and will show

a typical −3dB bandwidth of 240MHz. When the phase margin

is 64°, the closed-loop bandwidth is approximately √2

greater

than the value predicted by dividing GBP by the noise gain.