Datasheet

OPA657
12
SBOS197E
www.ti.com
To choose the values for both C
S
and C
F
, two parameters and
only three equations need to be solved. The first parameter is
the target high-frequency noise gain NG
2
, which should be
greater than the minimum stable gain for the OPA657. Here,
a target NG
2
of 10.5 will be used. The second parameter is the
desired low-frequency signal gain, which also sets the low-
frequency noise gain NG
1
. To simplify this discussion, we will
target a maximally flat 2nd-order low-pass Butterworth fre-
quency response (Q = 0.707). The signal gain of –2 shown in
Figure 4 will set the low frequency noise gain to
NG
1
= 1 + R
F
/R
G
(= 3 in this example). Then, using only these
two gains and the GBP for the OPA657 (1600MHz), the key
frequency in the compensation can be determined as:
Z
GBP
NG
NG
NG
NG
NG
O
=














1
2
1
2
1
2
112–––
Physically, this Z
0
(10.6MHz for the values shown above) is
set by 1/(2π • R
F
(C
F
+ C
S
)) and is the frequency at which the
rising portion of the noise gain would intersect unity gain if
projected back to 0dB gain. The actual zero in the noise gain
occurs at NG
1
• Z
0
and the pole in the noise gain occurs at
NG
2
• Z
0
. Since GBP is expressed in Hz, multiply Z
0
by 2π
and use this to get C
F
by solving:
C
RZNG
F
F
O
=
1
2
2
π •
(= 2.86pF)
Finally, since C
S
and C
F
set the high-frequency noise gain,
determine C
S
by [Using NG
2
= 10.5]:
C
S
= (NG
2
– 1)C
F
(= 27.2pF)
The resulting closed-loop bandwidth will be approximately
equal to:
f Z GBP
dB–3
0
≅
(= 130MHz)
For the values shown in Figure 4, the f
–3dB
will be approximately
130MHz. This is less than that predicted by simply dividing the
GBP product by NG
1
. The compensation network controls the
bandwidth to a lower value while providing the full slew rate at
the output and an exceptional distortion performance due to
increased loop gain at frequencies below NG
1
• Z
0
. The
capacitor values shown in Figure 4 are calculated for NG
1
= 3
and NG
2
= 10.5 with no adjustment for parasitics.
Figure 5 shows the measured frequency response for the
circuit of Figure 4. This is showing the expected gain of –2
with exceptional flatness through 70MHz and a –3dB band-
width of 170MHz.
The real benefit to this compensation is to allow a high slew
rate, exceptional DC precision op amp to provide a low
overshoot, fast settling pulse response. For a 1V output step,
the 700V/µs slew rate of the OPA657 will allow a rise time
limited edge rate (2ns for a 170MHz bandwidth). While unity-
gain stable op amps may offer comparable bandwidths, their
lower slew rates will extend the settling time for larger steps.
For instance, the OPA656 can also provide a 150MHz gain of
–2 bandwidth implying a 2.3ns transition time. However, the
lower slew rate of this unity gain stable amplifier (290V/µs) will
limit a 1V step transition to 3.5ns and delay the settling time as
the slewing transition is recovered. The combination of higher
slew rate and exceptional DC precision for the OPA657 can
yield one of the fastest, most precise, pulse amplifiers using
the circuit of Figure 4.
An added benefit to the compensation of Figure 4 is to
increase the loop gain above that achievable at comparable
gains by internally compensated amplifiers. The circuit of
Figure 4 will have lower harmonic distortion through 10MHz
than the OPA656 operated at a gain of –2.
R
F
500Ω
C
S
27pF
OPA657
+5V
–5V
V
O
= –2 • V
I
V
I
C
F
2.9pF
R
G
250Ω
12
9
6
3
0
–3
–6
–9
–12
–15
–18
Frequency (MHz)
Gain (3dB/div)
1 10 100 500
170MHz
FIGURE 4. Broadband Low Gain Inverting External Com-
pensation.
FIGURE 5. G = –2 Frequency Response with External
Compensation.