Datasheet

OPA657
13
SBOS197E
www.ti.com
DESIGN-IN TOOLS
DEMONSTRATION FIXTURES
Two printed circuit boards (PCBs) are available to assist in
the initial evaluation of circuit performance using the OPA657
in its two package options. Both of these are offered free of
charge as unpopulated PCBs, delivered with a user's guide.
The summary information for these fixtures is shown in
Table I.
The total output spot noise voltage can be computed as the
square root of the squared contributing terms to the output
noise voltage. This computation is adding all the contributing
noise powers at the output by superposition, then taking the
square root to get back to a spot noise voltage. Equation 1
shows the general form for this output noise voltage using
the terms shown in Figure 7:
(1)
E E I R kTR NG I R kTR NG
O
NI BN
SS
BI F F
=+
( )
+




+
( )
+
2
2
2
2
44
Dividing this expression by the noise gain (G
N
= 1 + R
F
/R
G
)
will give the equivalent input referred spot noise voltage at
the non-inverting input as shown in Equation 2:
(2)
E E I R kTR
IR
NG
kTR
NG
NNIBN
SS
BI F F
=+
( )
++






+
2
2
2
4
4
Putting high resistor values into Equation 2 can quickly
dominate the total equivalent input referred noise. A source
impedance on the noninverting input of 1.6kΩ will add a
Johnson voltage noise term equal to just that for the amplifier
itself (5nV/
Hz
). While the JFET input of the OPA657 is ideal
for high source impedance applications, both the overall
bandwidth and noise may be limited by these higher source
impedances in the non-inverting configuration of Figure 1.
FREQUENCY RESPONSE CONTROL
Voltage-feedback op amps exhibit decreasing closed-loop
bandwidth as the signal gain is increased. In theory, this
relationship is described by the Gain Bandwidth Product
(GBP) shown in the specifications. Ideally, dividing GBP by
the non-inverting signal gain (also called the Noise Gain, or
NG) will predict the closed-loop bandwidth. In practice, this
only holds true when the phase margin approaches 90°, as
it does in high-gain configurations. At low gains (increased
feedback factors), most high-speed amplifiers will exhibit a
more complex response with lower phase margin. The
OPA657 is compensated to give a maximally flat 2nd-order
Butterworth closed-loop response at a noninverting gain of
+10 (Figure 1). This results in a typical gain of +10 bandwidth
of 275MHz, far exceeding that predicted by dividing the
1600MHz GBP by 10. Increasing the gain will cause the
phase margin to approach 90° and the bandwidth to more
closely approach the predicted value of (GBP/NG). At a gain
of +50 the OPA657 will show the 32MHz bandwidth predicted
using the simple formula and the typical GBP of 1600MHz.
Inverting operation offers some interesting opportunities to
increase the available gain-bandwidth product. When the
source impedance is matched by the gain resistor (Figure 2),
The demonstration fixtures can be requested at the Texas
Instruments web site (www.ti.com) through the OPA657
product folder.
OPERATING SUGGESTIONS
SETTING RESISTOR VALUES TO MINIMIZE NOISE
The OPA657 provides a very low input noise voltage while
requiring a low 14mA of quiescent current. To take full advan-
tage of this low input noise, a careful attention to the other
possible noise contributors is required. Figure 6 shows the op
amp noise analysis model with all the noise terms included. In
this model, all the noise terms are taken to be noise voltage or
current density terms in either nV/
Hz
or pA/
Hz
.
4kT
R
G
R
G
R
F
R
S
OPA657
I
BI
E
O
I
BN
4kT = 1.6E –20J
at 290°K
E
RS
E
NI
√4kTR
S
√4kTR
F
*
*
*
FIGURE 6. Op Amp Noise Analysis Model.
ORDERING LITERATURE
PRODUCT PACKAGE NUMBER NUMBER
OPA657U SO-8 DEM-OPA-SO-1A SBOU009
OPA657N SOT23-5 DEM-OPA-SOT-1A SBOU010
TABLE I. Demonstration Fixtures by Package.