Datasheet
OPA657
11
SBOS197E
www.ti.com
WIDEBAND, HIGH SENSITIVITY, TRANSIMPEDANCE
DESIGN
The high GBP and low input voltage and current noise for the
OPA657 make it an ideal wideband-transimpedance ampli-
fier for moderate to high transimpedance gains. Unity-gain
stability in the op amp is not required for application as a
transimpedance amplifier. One transimpedance design ex-
ample is shown on the front page of the data sheet. Designs
that require high bandwidth from a large area detector with
relatively high transimpedance gain will benefit from the low
input voltage noise for the OPA657. This input voltage noise
is peaked up over frequency by the diode source capaci-
tance, and can, in many cases, become the limiting factor to
input sensitivity. The key elements to the design are the
expected diode capacitance (C
D
) with the reverse bias volt-
age (–V
B
) applied, the desired transimpedance gain, R
F
, and
the GBP for the OPA657 (1600MHz). Figure 3 shows a
design from a 50pF source capacitance diode through a
200kΩ transimpedance gain. With these three variables set
(and including the parasitic input capacitance for the OPA657
added to C
D
), the feedback capacitor value (C
F
) may be set
to control the frequency response.
This will give an approximate –3dB bandwidth set by:
f GBP R C Hz
dB F D−
=
3
2/)π
The example of Figure 3 will give approximately 5MHz flat
bandwidth using the 0.2pF feedback compensation.
If the total output noise is bandlimited to a frequency less
than the feedback pole frequency, a very simple expression
for the equivalent input noise current can be derived as:
II
kT
R
E
R
ECF
EQ
N
F
N
F
ND
=+ +
+
(
)
2
2
2
4
2
3
π
Where:
I
EQ
= Equivalent input noise current if the output noise is
bandlimited to F < 1/(2πR
F
C
F
).
I
N
= Input current noise for the op amp inverting input.
E
N
= Input voltage noise for the op amp.
C
D
= Diode capacitance.
F = Bandlimiting frequency in Hz (usually a postfilter prior
to further signal processing).
4kT = 1.6E – 21J at T = 290°K
Evaluating this expression up to the feedback pole frequency
at 3.9MHz for the circuit of Figure 3, gives an equivalent input
noise current of 3.4pA/
Hz
. This is much higher than the
1.2fA/
Hz
for just the op amp itself. This result is being
dominated by the last term in the equivalent input noise
expression. It is essential in this case to use a low voltage
noise op amp like the OPA657. If lower transimpedance gain,
wider bandwidth solutions are needed, consider the bipolar
input OPA686 or OPA687. These parts offer comparable
gain bandwidth products but much lower input noise voltage
at the expense of higher input current noise.
LOW GAIN COMPENSATION
Where a low gain is desired, and inverting operation is
acceptable, a new external compensation technique may be
used to retain the full slew rate and noise benefits of the
OPA657 while maintaining the increased loop gain and the
associated improvement in distortion offered by the decom-
pensated architecture. This technique shapes the loop gain
for good stability while giving an easily controlled 2nd-order
low-pass frequency response. Considering only the noise
gain for the circuit of Figure 4, the low-frequency noise gain,
(N
G1
) will be set by the resistor ratios while the high fre-
quency noise gain (N
G2
) will be set by the capacitor ratios.
The capacitor values set both the transition frequencies and
the high-frequency noise gain. If this noise gain, determined
by N
G2
= 1 + C
S
/C
F
, is set to a value greater than the
recommended minimum stable gain for the op amp and the
noise gain pole, set by 1/R
F
C
F
, is placed correctly, a very well
controlled 2nd-order low-pass frequency response will result.
To achieve a maximally flat 2nd-order Butterworth frequency
response, the feedback pole should be set to:
12 4/( ) ( /( ))ππRC GBP RC
FF FD
=
Adding the common-mode and differential mode input capaci-
tance (0.7 + 4.5)pF to the 50pF diode source capacitance of
Figure 3, and targeting a 200kΩ transimpedance gain using
the 1600MHz GBP for the OPA657 will require a feedback
pole set to 3.5MHz. This will require a total feedback capaci-
tance of 0.2pF. Typical surface-mount resistors have a para-
sitic capacitance of 0.2pF, therefore, while Figure 3 shows a
0.2pF feedback-compensation capacitor, this will actually be
the parasitic capacitance of the 200kΩ resistor.
R
F
200kΩ
Supply Decoupling
Not Shown
C
D
50pF
λ
OPA657
+5V
–5V
–V
B
I
D
V
O
=
I
D
R
F
C
F
0.2pF
FIGURE 3. Wideband, Low Noise, Transimpedance Amplifier.