Datasheet

OPA656
11
SBOS196G
www.ti.com
is peaked up over frequency by the diode source capaci-
tance, and can, in many cases, become the limiting factor to
input sensitivity. The key elements to the design are the
expected diode capacitance (C
D
) with the reverse bias volt-
age (–V
B
) applied, the desired transimpedance gain, R
F
, and
the GBP for the OPA656 (230MHz). Figure 3 shows a design
from a 25pF source capacitance diode through a 50kΩ
transimpedance gain. With these 3 variables set (including
the parasitic input capacitance for the OPA656 added to C
D
),
the feedback capacitor value (C
F
) may be set to control the
frequency response.
If the total output noise is bandlimited to a frequency less
than the feedback pole frequency (1/R
F
C
F
), a very simple
expression for the equivalent input noise current can be
derived as:
II
kT
R
E
R
ECF
EQ
N
F
N
F
ND
=+ +






+
( )
2
2
2
4
2
3
π
Where:
i
EQ
= Equivalent input noise current if the output noise is
bandlimited to F < 1/(2πR
F
C
D
).
i
N
= Input current noise for the op amp inverting input.
e
N
= Input voltage noise for the op amp.
C
D
= Diode capacitance.
F = Bandlimiting frequency in Hz (usually a postfilter prior
to further signal processing).
4kT = 1.6E – 20J at 290°K.
Evaluating this expression up to the feedback pole frequency
at 3.8MHz for the circuit of Figure 3, gives an equivalent input
noise current of 2.7pA/
Hz
. This is much higher than the
1.3fA/
Hz
for just the op amp itself. This result is being
dominated by the last term in the equivalent input noise
current expression. It is essential in this case to use a low
voltage noise op amp.
DESIGN-IN TOOLS
DEMONSTRATION FIXTURES
Two printed circuit boards (PCBs) are available to assist in
the initial evaluation of circuit performance using the OPA656
in its two package options. Both of these are offered free of
charge as unpopulated PCBs, delivered with a user's guide.
The summary information for these fixtures is shown in
Table I.
To achieve a maximally flat 2nd-order Butterworth frequency
response, the feedback pole should be set to:
12 4/( ) ( /( ))ππRC GBP RC
FF FD
=
Adding the common mode and differential mode input ca-
pacitance (0.7 + 2.8)pF to the 25pF diode source capaci-
tance of Figure 3, and targeting a 50kΩ transimpedance gain
using the 230MHz GBP for the OPA656 will require a
feedback pole set to 3.8MHz. This will require a total feed-
back capacitance of 0.8pF. Typical surface-mount resistors
have a parasitic capacitance of 0.2pF leaving the required
0.6pF value shown in Figure 3 to get the required feedback
pole.
This will give an approximate –3dB bandwidth set by:
f GBP R C Hz
dB F D−
=
3
2/)π
The example of Figure 3 will give approximately 5.7MHz flat
bandwidth using the 0.6pF feedback compensation.
FIGURE 3. Wideband, Low-Noise, Transimpedance Amplifier.
R
F
50kΩ
Supply Decoupling
Not Shown
C
D
25pF
λ
OPA656
+5V
–5V
–V
B
I
D
V
O
= I
D
R
F
C
F
0.6pF
ORDERING LITERATURE
PRODUCT PACKAGE NUMBER NUMBER
OPA656U SO-8 DEM-OPA-SO-1A SBOU009
OPA656N SOT23-5 DEM-OPA-SOT-1A SBOU010
TABLE I. Demonstration Fixtures by Package.
The demonstration fixtures can be requested at the Texas
Instruments web site (www.ti.com) through the OPA656
product folder.