Datasheet

MAX1778/MAX1880–MAX1885
Quad-Output TFT LCD DC/DC
Converters with Buffer
25
Maxim Integrated
where I
MAIN
includes the primary load current and the
input supply currents for the charge pumps (see the
Charge-Pump Input Power and Efficiency
Considerations section), linear regulator, and VCOM
buffer.
The linear regulator generates an output voltage by dis-
sipating power across an internal pass transistor, so
the power dissipation is simply the load current times
the input-to-output voltage differential:
When driving an external transistor, the internal linear
regulator provides the base drive current. Depending
on the external transistor’s current gain (β) and the
maximum load current, the power dissipated by the
internal linear regulator can still be significant:
The charge pumps provide regulated output voltages
by dissipating power in the low-side n-channel
MOSFET, so they could be modeled as linear regula-
tors followed by unregulated charge pumps. Therefore,
their power dissipation is similar to a linear regulator:
where N is the number of charge-pump stages, V
DIODE
is the diodes’ forward voltage, and V
SUPD
is the
positive charge-pump diode supply (Figure 4).
The VCOM buffer’s power dissipation depends on the
capacitive load (C
LOAD
) being driven, the peak-to-
peak voltage change (V
P-P
) across the load, and the
load’s switching rate:
To find the total power dissipated in the device, the
power dissipated by each regulator and the buffer must
be added together:
The maximum allowed power dissipation is 975mW (24-
pin TSSOP)/879mW (20-pin TSSOP) or:
P
MAX
= (T
J(MAX
) - T
A
)/(θ
JB
+ θ
BA
)
where T
J
- T
A
is the temperature difference between
the controller’s junction and the surrounding air, θ
JB
(or
θ
JC
) is the thermal resistance of the package to the
board, and θ
BA
is the thermal resistance from the PCB
to the surrounding air.
Design Procedure
Main Step-Up Converter
Output-Voltage Selection
Adjust the output voltage by connecting a voltage-
divider from the output (V
MAIN
) to FB to GND (see the
Typical Operating Circuit ). Select R2 in the 10kΩ to
50kΩ range. Calculate R1 with the following equations:
R1 = R2 [(V
MAIN
/V
REF
) - 1]
where V
REF
= 1.25V. V
MAIN
can range from V
IN
to 13V.
Inductor Selection
Inductor selection depends upon the minimum required
inductance value, saturation rating, series resistance, and
size. These factors influence the converter’s efficiency,
maximum output load capability, transient response time,
and output-voltage ripple. For most applications, values
between 4.7µH and 22µH work best with the controller’s
switching frequency (Tables 1 and 2).
The inductor value depends on the maximum output
load the application must support, input voltage, output
voltage, and switching frequency. With high inductor
values, the MAX1778/MAX1880–MAX1885 source high-
er output currents, have less output ripple, and enter
continuous conduction operation with lighter loads;
however, the circuit’s transient response time is slower.
On the other hand, low-value inductors respond faster
to transients, remain in discontinuous conduction oper-
ation, and typically offer smaller physical size for a
given series resistance and current rating. The equa-
tions provided here include a constant LIR, which is the
ratio of the peak-to-peak AC inductor current to the
average DC inductor current. For a good compromise
between the size of the inductor, power loss, and
output-voltage ripple, select an LIR of 0.3 to 0.5. The
inductance value is then given by:
L
V
V
VV
I f LIR
MIN
IN MIN
MAIN
MAIN IN MIN
MAIN MAX OSC
() ()
()
=
⎛
⎝
⎜
⎞
⎠
⎟
⎛
⎝
⎜
⎞
⎠
⎟
⎛
⎝
⎜
⎞
⎠
⎟
2
1
-
η
PP P
PPP
TOTAL STEP UP LDO INT
NEG POS BUF
()
=+
+++
-
PVCfV
BUF P P LOAD LOAD SUPB
=
-
PIV VNV
PIV VNV V
NEG NEG SUPN DIODE NEG
POS POS SUPP DIODE SUPD POS
=
()
[]
=
()
+
[]
--
--
2
2
P
I
VV V
IVV
LDO INT
LDO
SUPL LDO
LDOOUT SUPL LDOOUT
()
.
( )
=+
()
[]
=
β
-
-
07
PIVV
LDO INT LDO SUPL LDO()
( )= -