Datasheet

LTC3851-1
17
38511fa
APPLICATIONS INFORMATION
is low. When the topside MOSFET is to be turned on, the
driver places the C
B
voltage across the gate source of the
MOSFET. This enhances the MOSFET and turns on the
topside switch. The switch node voltage, SW, rises to V
IN
and the BOOST pin follows. With the topside MOSFET on,
the boost voltage is above the input supply:
V
BOOST
= V
IN
+ V
INTVCC
The value of the boost capacitor C
B
needs to be 100 times
that of the total input capa citance of the topside MOSFET.
The reverse break down of the external Schottky diode
must be greater than V
IN(MAX)
.
Undervoltage Lockout
The LTC3851-1 has two functions that help protect the
controller in case of undervoltage conditions. A precision
UVLO comparator constantly monitors the INTV
CC
voltage
to ensure that an adequate gate-drive voltage is present.
It locks out the switching action when INTV
CC
is below
3.2V. To prevent oscillation when there is a disturbance
on the INTV
CC
, the UVLO comparator has 400mV of preci-
sion hysteresis.
Another way to detect an undervoltage condition is to
monitor the V
IN
supply. Because the RUN pin has a precision
turn-on reference of 1.25V, one can use a resistor divider
to V
IN
to turn on the IC when V
IN
is high enough.
C
IN
Selection
In continuous mode, the source current of the top N-channel
MOSFET is a square wave of duty cycle V
OUT
/V
IN
. To
prevent large voltage transients, a low ESR input capacitor
sized for the maximum RMS current must be used. The
maximum RMS capacitor current is given by:
II
V
V
V
V
RMS O MAX
OUT
IN
IN
OUT
≅
⎛
⎝
⎜
⎞
⎠
⎟
()
/
–1
12
This formula has a maximum at V
IN
= 2V
OUT
, where I
RMS
=
I
O(MAX)
/2. This simple worst-case condition is com monly
used for design because even signifi cant deviations do not
offer much relief. Note that capacitor manufacturers’ ripple
current ratings are often based on only 2000 hours of life.
This makes it advisable to further derate the capacitor or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may also be paralleled to meet
size or height requirements in the design. Always consult
the manufacturer if there is any question.
C
OUT
Selection
The selection of C
OUT
is primarily determined by the
effective series resistance, ESR, to minimize voltage
ripple. The output ripple, ∆V
OUT
, in continuous mode is
determined by:
∆∆V I ESR
fC
OUT L
OUT
≈ +
⎛
⎝
⎜
⎞
⎠
⎟
1
8
where f = operating frequency, C
OUT
= output capaci tance
and ∆I
L
= ripple current in the inductor. The output ripple is
highest at maximum input voltage since ∆I
L
increases with
input voltage. Typically, once the ESR requirement for C
OUT
has been met, the RMS current rating generally far exceeds
the I
RIPPLE(P-P)
requirement. With ∆I
L
= 0.3I
OUT(MAX)
and
allowing 2/3 of the ripple to be due to ESR, the output
ripple will be less than 50mV at maximum V
IN
and:
C Required ESR < 2.2R
OUT SENSE
C
fR
OUT
SENSE
>
1
8
The fi rst condition relates to the ripple current into the ESR
of the output capacitance while the second term guaran tees
that the output capacitance does not signifi cantly discharge
during the operating frequency period due to ripple current.
The choice of using smaller output capaci tance increases
the ripple voltage due to the discharging term but can be
compensated for by using capacitors of very low ESR to
maintain the ripple voltage at or below 50mV. The I
TH
pin
OPTI-LOOP compensation compo nents can be optimized
to provide stable, high perfor mance transient response
regardless of the output capaci tors selected.
The selection of output capacitors for applications with
large load current transients is primarily determined by the
voltage tolerance specifi cations of the load. The resistive
component of the capacitor, ESR, multiplied by the load
current change, plus any output voltage ripple must be
within the voltage tolerance of the load.