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LTC1702

1702fa

Dual 550kHz Synchronous

2-Phase Switching Regulator Controller

The LTC

1702 is a dual switching regulator controller opti-

mized for high efficiency with low input voltages. It includes

two complete, on-chip, independent switching regulator

controllers each designed to drive a pair of external N-channel

MOSFET devices in a voltage mode feedback, synchronous

buck configuration. The LTC1702 uses a constant-frequency,

true PWM design switching at 550kHz, minimizing external

component size and cost and maximizing load transient

performance. The synchronous buck architecture automati-

cally shifts to discontinuous and then to Burst Mode

operation as the output load decreases, ensuring maximum

efficiency over a wide range of load currents.

The LTC1702 features an onboard reference trimmed to

0.5% and can provide better than 1% regulation at the

converter outputs. Open-drain logic outputs indicate whether

either output has risen to within 5% of the final output voltage

and an optional latching FAULT mode protects the load if the

output rises 15% above the intended voltage. Each channel

can be enabled independently; with both channels disabled,

the LTC1702 shuts down and supply current drops below

100µA.

■

Two Independent Controllers in One Package

■

Two Sides Run Out-of-Phase to Minimize C

■

All N-Channel External MOSFET Architecture

■

No External Current Sense Resistors

■

Excellent Output Regulation: 1% Total Output

Accuracy

■

550kHz Switching Frequency Minimizes External

Component Size

■

1A to 25A Output Current per Channel

■

High Efficiency over Wide Load Current Range

■

Quiescent Current Drops Below 100µA in Shutdown

■

Small 24-Pin Narrow SSOP Package

■

Microprocessor Core and I/O Supplies

■

Multiple Logic Supply Generator

■

Distributed Power Applications

■

High Efficiency Power Conversion

Dual Output High Power 3.3V/2.5V Logic Supply

DESCRIPTIO

FEATURES

APPLICATIO S

TYPICAL APPLICATIO

LTC1702

BOOST1

BG1

TG1

SW1

MAX1

PGOOD1

FCB

RUN/SS

COMP1

SGND

FB1

MAX2

BOOST2

BG2

TG2

SW2

PGND

PGOOD2

FAULT

RUN/SS2

COMP2

FB2

1µF

10Ω

1µF

27k

1.6k

68k

1702 TA01

1.2k

10k

OUT1

180µF

×4

820pF

27pF

680pF

47k

1µF

OUT1

2.5V

AT 15A

1µF

4.75k

Q2Q1

Q4Q3

OUT1

, C

OUT2

: PANASONIC EEFUE0G181R

: KEMET TS10X337M010AS

D1, D2: MOTOROLA MBR0520LT1

D3, D4: MOTOROLA MBRS320T3

L1, L2: SUMIDA CEP125-1R0

Q1 TO Q8: FAIRCHILD FDS6670A

PWRGD1

1µH

10µF

1µF

= 5V ±10%

1µF

330µF

×3

680pF

27pF

3300pF

15.8k

4.99k

10k

1µH

OUT2

180µF

×4

1µF

OUT2

3.3V/15A

PWRGD2

FAULT

10k

, LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode

is a registered trademark of Linear Technology Corporation. All other trademarks are the

property of their respective owners. Protected by U.S. Patents including 5481178,

5847554, 6304066, 6307356, 6580258.

Summary of content (36 pages)

PAGE 1
LTC1702 Dual 550kHz Synchronous 2-Phase Switching Regulator Controller U FEATURES DESCRIPTIO ■ The LTC®1702 is a dual switching regulator controller optimized for high efficiency with low input voltages. It includes two complete, on-chip, independent switching regulator controllers each designed to drive a pair of external N-channel MOSFET devices in a voltage mode feedback, synchronous buck configuration.
PAGE 2
LTC1702 U W U U W W W ABSOLUTE MAXIMUM RATINGS PACKAGE/ORDER INFORMATION (Note 1) TOP VIEW Supply Voltage VCC ........................................................................ 7V BOOSTn ............................................................... 15V BOOSTn – SWn .................................................... 7V Input Voltage SWn .......................................................... – 1V to 8V All Other Inputs ......................... – 0.3V to VCC + 0.
PAGE 3
LTC1702 ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C. VCC = 5V unless otherwise specified.
PAGE 4
LTC1702 U W TYPICAL PERFOR A CE CHARACTERISTICS Efficiency vs Load Current VIN = 5V VOUT = 2.5V 90 35 VIN = 5V VOUT = 1.8V ILOAD = 0A-10A-0A ±2.2% MAX DEVIATION VOUT = 3.3V VOUT = 1.
PAGE 5
LTC1702 U U U PIN FUNCTIONS PVCC (Pin 1): Driver Power Supply Input. PVCC provides power to the two BGn output drivers. PVCC must be connected to a voltage high enough to fully turn on the external MOSFETs QB1 and QB2. PVCC should generally be connected directly to VIN. PVCC requires at least a 1µF bypass capacitor directly to PGND. BOOST1 (Pin 2): Controller 1 Top Gate Driver Supply. The BOOST1 pin supplies power to the floating TG1 driver. BOOST1 should be bypassed to SW1 with a 1µF capacitor.
PAGE 6
LTC1702 U U U PIN FUNCTIONS FB2 (Pin 14): Controller 2 Feedback Input. See FB1. COMP2 (Pin 15): Controller 2 Loop Compensation. See COMP1. RUN/SS2 (Pin 16): Controller 2 Run/Soft-start. See RUN/ SS1. FAULT (Pin 17): Output Overvoltage Fault (Latched). The FAULT pin is an open-drain output with an internal 10µA pull-up. If either regulated output voltage rises more than 15% above its programmed value for more than 25µs, the FAULT output will go high and the entire LTC1702 will be disabled.
PAGE 7
LTC1702 TEST CIRCUIT Test Circuit 1 5V 0.1µF IBOOST1 ICC VCC BOOST1 fOSC MEASURED 5V CL CL 10k IBOOST2 TG2 BG1 BG2 SW1 SW2 IMAX1 IMAX2 CL LTC1702 PGOOD2 RUN/SS1 RUN/SS2 5V VPGOOD2 VFAULT 2k COMP2 FB1 FB2 GND CL 10k PGOOD1 COMP1 VFB1 100µF BOOST2 FAULT 2k + PVCC TG1 FCB VPGOOD1 IPVCC VFB2 PGND 1702 TC U W U U APPLICATIONS INFORMATION OVERVIEW The LTC1702 is a dual, step-down (buck), voltage mode feedback switching regulator controller.
PAGE 8
LTC1702 U W U U APPLICATIONS INFORMATION 2-Step Conversion “2-step” architectures use a primary regulator to convert the input power source (batteries or AC line voltage) to an intermediate supply voltage, often 5V. This intermediate voltage is then converted to the low voltage, high current supplies required by the system using a secondary regulator— the LTC1702.
PAGE 9
LTC1702 U W U U APPLICATIONS INFORMATION remember that the power losses are proportional to IRMS2, meaning that the actual power wasted is reduced by a factor of 3.75. The reduced input ripple voltage also means less power is lost in the input power path, which could include batteries, switches, trace/connector resistances and protection circuitry. Improvements in both conducted and radiated EMI also directly accrue as a result of the reduced RMS input current and voltage.
PAGE 10
LTC1702 U W U U APPLICATIONS INFORMATION amp of continuous current with peak currents up to 5A to slew large MOSFET gates quickly. The external MOSFETs are connected with the drain of QT attached to the input supply and the source of QT at the switching node SW. QB is the synchronous rectifier with its drain at SW and its source at PGND. SW is connected to one end of the inductor, with the other end connected to VOUT. The output capacitor is connected from VOUT to PGND.
PAGE 11
LTC1702 U U W U APPLICATIONS INFORMATION improves loop phase margin (see Figure 3). The Feedback Loop/Compensation section contains a detailed explanation of type 3 feedback loops. + COMP 0.8V FB R1 FB – C3 R3 VOUT RB C2 R2 C1 1702 F03 Figure 3. “Type 3” Feedback Loop MIN/MAX Two additional feedback loops keep an eye on the primary feedback amplifier and step in if the feedback node moves ±5% from its nominal 800mV value.
PAGE 12
LTC1702 U W U U APPLICATIONS INFORMATION SHUTDOWN/SOFT-START Each half of the LTC1702 has a RUN/SS pin. The RUN/SS pins perform two functions: when pulled to ground, each shuts down its half of the LTC1702, and each acts as a conventional soft-start pin, enforcing a maximum duty cycle limit proportional to the voltage at RUN/SS. An internal 3.5µA current source pull-up is connected to each RUN/SS pin, allowing a soft-start ramp to be generated with a single external capacitor to ground. The 3.
PAGE 13
LTC1702 U W U U APPLICATIONS INFORMATION Any time QB is on and the current flowing to the output is reasonably large, the SW node at the drain of QB will be somewhat negative with respect to PGND. The LTC1702 senses this voltage and inverts it to allow it to compare the sensed voltage with a positive voltage at the IMAX pin. The IMAX pin includes a trimmed 10µA pull-up, enabling the user to set the voltage at IMAX with a single resistor, RIMAX, to ground.
PAGE 14
LTC1702 U W U U APPLICATIONS INFORMATION ripple voltage and frequency remain constant at all loads, easing filtering requirements. Circuits that take advantage of this behavior can force the LTC1702 to operate in continuous mode at all loads by tying the FCB (Force Continuous Bar) pin to ground. Discontinuous Mode To minimize the efficiency loss due to reverse current flow at light loads, the LTC1702 switches to a second mode of operation: discontinuous mode (Figure 5b).
PAGE 15
LTC1702 U W U U APPLICATIONS INFORMATION Once again, the LTC1702 switches to a new mode to minimize efficiency loss: Burst Mode operation. As the circuit goes deeper and deeper into discontinuous mode, the total time QT and QB are on reduces. However, the ratio of the time that QT is on to the time that QB is on must remain constant for the output to stay in regulation. An internal timer circuit forces QT to stay on for at least 10% of a normal switching cycle.
PAGE 16
LTC1702 U W U U APPLICATIONS INFORMATION below ground. If latched FAULT mode is used, a Schottky diode should be added with its cathode at the output and its anode at ground to clamp the negative voltage to a safe level and prevent possible damage to the load and the output capacitors. Note that in overvoltage conditions, the MAX comparator will kick in at just +5%, turning QB on continuously long before the output reaches +15%.
PAGE 17
LTC1702 U W U U APPLICATIONS INFORMATION QB). This capacitance is composed of MOSFET channel charge, actual parasitic drain-source capacitance and Miller-multiplied gate-drain capacitance, but can be approximated as a single capacitance from gate to source. Regardless of where the charge is going, the fact remains that it all has to come out of VCC to turn the MOSFET gate on, and when the MOSFET is turned back off, that charge all ends up at ground.
PAGE 18
LTC1702 U W U U APPLICATIONS INFORMATION bypass capacitor gets exercised in three ways: its ESR must be low enough to keep the initial drop as QT turns on within reason (100mV or so); its RMS current capability must be adequate to withstand the 4.6ARMS ripple current at the input and the capacitance must be large enough to maintain the input voltage until the input supply can make up the difference.
PAGE 19
LTC1702 U U W U APPLICATIONS INFORMATION from CIN (time point A). 50% of the way through, TG2 turns on and the total current is 13A (time point B). Shortly thereafter, TG1 turns off and the current drops to 10A (time point C). Finally, TG2 turns off and the current spends a short time at 0 before TG1 turns on again (time point D). ( ) ( ) (10A • 0.16) + (0A • 0.18) = 5.18A IAVG = 3A • 0.5 + 13A • 0.16 + Now we can calculate the RMS current.
PAGE 20
LTC1702 U W U U APPLICATIONS INFORMATION tantalum capacitors specifically designed for use with switching regulators. When choosing a tantalum input capacitor, make sure that it is rated to carry the RMS current that the LTC1702 will draw. If the data sheet doesn’t give an RMS current rating, chances are the capacitor isn’t surge tested. Don’t use it! OUTPUT BYPASS CAPACITOR The output bypass capacitor has quite different requirements from the input capacitor.
PAGE 21
LTC1702 U U W U APPLICATIONS INFORMATION feedback amplifier, on the other hand, gives us a handle with which to adjust the AC response. The goal is to have 180° phase shift at DC (so the loop regulates) and something less than 360° phase shift at the point that the loop gain falls to 0dB. The simplest strategy is to set up the feedback amplifier as an inverting integrator, with the 0dB frequency lower than the LC pole (Figure 9).
PAGE 22
LTC1702 U U W U APPLICATIONS INFORMATION “Type 3” loops (Figure 11) use two poles and two zeros to obtain a 180° phase boost in the middle of the frequency band. A properly designed type 3 circuit can maintain acceptable loop stability even when low output capacitor ESR causes the LC section to approach 180° phase shift well above the initial LC roll-off. As with a type 2 circuit, the loop should cross through 0dB in the middle of the phase bump to maximize phase margin.
PAGE 23
LTC1702 U W U U APPLICATIONS INFORMATION nodes don’t corrupt the measurements or damage the analyzer. Finally, choose a convenient resistor value for R1 (10k is usually a good value). Now calculate the remaining values: If breadboard measurement is not practical, a SPICE simulation can be used to generate approximate gain/ phase curves. Plug the expected capacitor, inductor and MOSFET values into the following SPICE deck and generate an AC plot of V(VOUT)/V(COMP) in dB and phase of V(OUT) in degrees.
PAGE 24
LTC1702 U W U U APPLICATIONS INFORMATION CURRENT LIMIT PROGRAMMING Accuracy Trade-Offs Programming the current limit on the LTC1702 is straightforward. The IMAX pin sets the current limit by setting the maximum allowable voltage drop across QB (the bottom MOSFET) before the current limit circuit engages. The voltage across QB is set by its on-resistance and the current flowing in the inductor, which is the same as the output current.
PAGE 25
LTC1702 U W U U APPLICATIONS INFORMATION VIN VOUT(AUX) + + CIN TG COUT(AUX) QT LTC1702 FCB BG + QB VOUT COUT RFCB1 RFCB2 1702 F08 Figure 13. Regulating an Auxiliary Output with the FCB Pin with 10mV of hysteresis, allowing fairly precise control of the auxiliary voltage. If the LTC1702 is in discontinuous or Burst Mode operation and the auxiliary output voltage drops, the FCB pin will trip and the LTC1702 will resume continuous operation regardless of the load on the main output.
PAGE 26
LTC1702 U U W U APPLICATIONS INFORMATION to an intermediate supply voltage, often 5V. The LTC1702 then converts the intermediate voltage to the low voltage, high current supplies required by the system. Compared to a 1-step converter that converts a high input voltage directly to a very low output voltage, the 2-step converter exhibits superior transient response, smaller component size and equivalent efficiency. Thermal management and layout complexity are also improved with a 2-step approach.
PAGE 27
LTC1702 U W U U APPLICATIONS INFORMATION 2-Step Efficiency Calculation Maximizing High Load Current Efficiency Calculating the efficiency of a 2-step converter system involves some subtleties. Simply multiplying the efficiency of the primary 5V or 3.3V supply by the efficiency of the 1.8V or 1.5V supply underestimates the actual efficiency, since a significant fraction of the total power is drawn from the 3.3V and 5V rails in a typical system.
PAGE 28
LTC1702 U W U U APPLICATIONS INFORMATION Maximizing Low Load Current Efficiency Low load current efficiency depends strongly on proper operation in discontinuous and Burst Mode operations. In an ideally optimized system, discontinuous mode reduces conduction losses but not switching losses, since each power MOSFET still switches on and off once per cycle. In a typical system, there is additional loss in discontinuous mode due to a small amount of residual current left in the inductor when QB turns off.
PAGE 29
LTC1702 U U W U APPLICATIONS INFORMATION cycles. Over just a few cycles, however, the external components conspire to limit the speed that the output can move. Consider our typical 5V to 1.6V circuit, subjected to a 1A to 5A load transient. Initially, the loop is in regulation and the DC current in the output capacitor is zero. Suddenly, an extra 4A start flowing out of the output capacitor while the inductor is still supplying only 1A.
PAGE 30
LTC1702 U W U U APPLICATIONS INFORMATION Making L smaller than this optimum value yields little or no improvement in transient response. As the output voltage recovers, the inductor current will briefly rise above the level of the output current to replenish the charge lost from the output capacitor. With a properly compensated loop, the entire recovery time will be inside of 10µs.
PAGE 31
LTC1702 U W U U APPLICATIONS INFORMATION it goes out of spec. Similarly, at full load, the output current can only decrease, causing a positive shift in the output voltage; the initial low value allows it to rise further before the spec is exceeded. The primary benefit of voltage positioning is it increases the allowable ESR of the output capacitors, saving cost. An additional bonus is that at maximum load, the output voltage is near the minimum allowable, decreasing the power dissipated in the load.
PAGE 32
LTC1702 U W U U APPLICATIONS INFORMATION FAULT BEHAVIOR Changing the Output Voltage on the Fly Some applications use a switching scheme attached to the feedback resistors to allow the system to adjust the LTC1702 output voltage. The voltage can be changed on the fly if desired, but care must be taken to avoid tripping the overvoltage fault circuit.
PAGE 33
C11 330pF COUT1 470µF ×2 VOUT1 1.8V 10A + R21, 13k R11 10k 0.1% Q21 Q11 CSS1 0.1µF RIMAX1, 22k D2 MBR330T CCP1 1µF 1µF CIN, COUT1, COUT2: KEMET T510X477M006AS C21 680pF RB1 7.96k 0.1% C31 560pF R31, 4.7k LEXT1 1µH 12A DCP1 MBR0530T RUN/SS1 GND CCP2 1µF CSS2 0.
PAGE 34
100k 10k 10k 1% VIN 4.75k 0.1% 0.01µF 0.01µF 0.1µF PGOOD 10k 4.7k 0.1µF TG2 RUN/SS1 FAULT SGND FCB BG2 PGND BG1 IMAX2 SW2 IMAX1 PGOOD2 RUN/SS2 BOOST2 SW1 BOOST1 PGOOD1 COMP2 FB2 + 33k 33k 1µF D1 0.1µF 0.1µF 10Ω CIN2 100µF TG1 PVCC LTC1702 COMP1 FB1 VCC CVCC 10µF 10Ω LT1006 + 34 – 1µF 10k 10k D3 D2 CIN1 10µF ×2 QB1A QT1A 0.004Ω 0.5W QB1B QT1B 0.004Ω 0.
PAGE 35
LTC1702 U PACKAGE DESCRIPTION GN Package 24-Lead Plastic SSOP (Narrow .150 Inch) (Reference LTC DWG # 05-08-1641) .337 – .344* (8.560 – 8.738) 24 23 22 21 20 19 18 17 16 15 1413 .033 (0.838) REF .045 ±.005 .229 – .244 (5.817 – 6.198) .254 MIN .150 – .157** (3.810 – 3.988) .150 – .165 1 .0165 ± .0015 2 3 4 5 6 7 8 9 10 11 12 .0250 BSC RECOMMENDED SOLDER PAD LAYOUT .015 ± .004 × 45° (0.38 ± 0.10) .0075 – .0098 (0.19 – 0.25) .0532 – .0688 (1.35 – 1.75) .004 – .0098 (0.102 – 0.
PAGE 36
LTC1702 U TYPICAL APPLICATION Low Cost Dual Supply with 2.5V Keepalive VIN 5V ±10% IN + 10Ω 4.75k 0.1% 10k 0.1% + 10µF VCC OUT LT1761 16.2k 0.1% GND ADJ MBR0530T PVCC FB2 330pF 68k 1µF CIN 16.9k 0.1% BOOST2 56pF L2 1µF QT2 TG2 COMP2 VOUT2 2.5V/7A 2.45V/100mA STANDBY SW2 RUN/SS2 RUN/SS1 1k QSS1 QSS2 STBY/ON 0.1µF BG2 47k + QB2 COUT2 IMAX2 FAULT LTC1702 BOOST1 1k 20k 1% 56k 220pF MBR0530T 1µF QT1B FB1 TG1 39pF COMP1 QT1A BG1 FCB 16k 1% L1 VOUT1 1.