Datasheet
Data Sheet ADA4895-1/ADA4895-2
Rev. A | Page 21 of 24
WIDEBAND PHOTOMULTIPLIER PREAMPLIFIER
A decompensated amplifier can provide significantly greater
speed in transimpedance applications than a unity-gain stable
amplifier. The speed increases by the square root of the ratio of
the bandwidth of the two amplifiers; that is, a 1 GHz GBP amplifier
is 10 times faster than a 10 MHz amplifier in the same trans-
impedance application if all other parameters are kept constant.
Additionally, the input voltage noise normally dominates the
total output rms noise because it is multiplied by the capacitive
noise gain network.
( )
F
DFM
S
C
CCCC +++
In the case of the ADA4895-1/ADA4895-2, the input noise is
low, but the capacitive noise gain network must be kept greater
than 10 for stability reasons.
One disadvantage of using the ADA4895-1/ADA4895-2 in
transimpedance applications is that the input current and input
current noise can create large offsets and output voltage noise
when coupled with an excessively high feedback resistance. Despite
these two issues, the ADA4895-1/ADA4895-2 noise and gain
bandwidth can provide a significant increase in performance
within certain transimpedance ranges.
Figure 55 shows an I/V converter with an electrical model of a
photomultiplier.
–
+
V
OUT
V
B
C
F
+ C
S
C
D
C
M
C
M
R
F
R
SH
C
S
I
PHOTO
C
F
R
F
10186-050
Figure 55. Wideband Photomultiplier Preamplifier
The basic transfer function is
FF
F
PHOTO
OUT
RsC
RI
V
+
×
=
1
where I
PHOTO
is the output current of the photomultiplier, and
the parallel combination of R
F
and C
F
sets the signal bandwidth.
The stable bandwidth attainable with this preamplifier is a function
of R
F
, the gain bandwidth product of the amplifier, and the total
capacitance at the summing junction of the amplifier, including C
S
and the amplifier input capacitance.
R
F
and the total capacitance produce a pole in the loop trans-
mission of the amplifier that can result in peaking and instability.
Adding C
F
creates a zero in the loop transmission that compensates
for the pole effect and reduces the signal bandwidth. It can be
shown that the signal bandwidth resulting in a 45° phase margin
(f
(45)
) is defined as follows:
( )
S
F
45
CR
GBP
f
××
=
π2
where:
GBP is the gain bandwidth product.
R
F
is the feedback resistance.
C
S
is the total capacitance at the amplifier summing junction
(amplifier + photomultiplier + board parasitics).
The value of C
F
that produces f
(45)
is
GBPR
C
C
F
S
F
××
=
π2
The frequency response in this case shows approximately 2 dB
of peaking and 15% overshoot. Doubling C
F
and reducing the
bandwidth by half results in a flat frequency response with
approximately 5% transient overshoot.
The output noise over frequency for the preamplifier is shown
in Figure 56.
FREQUENCY (Hz)
VOLTAGE NOISE
RF NOISE
f
1
NOISE DUE TO AMPLIFIER
ven
f
2
1
2
π
R
F
f
1
=
f
2
=
1
2
π
R
F
C
F
f
3
=
GBP
ven (C
S
+ C
M
+ C
F
+ C
D
)
/C
F
f
3
10186-051
(C
S
+ C
M
+ C
F
+ C
D
)
(C
S
+ C
M
+ C
F
+ C
D
)
/C
F
(nV/ Hz)
Figure 56. Photomultiplier Voltage Noise Contributions
Table 12. RMS Noise Contributions of Photomultiplier
Preamplifier
Contributor Expression
R
F
5714 .fRkT
2
F
×××
Amplifier
ven
( )
57.1××
+++
×
3
F
DFM
S
f
C
CCCC
ven
Amplifier
ien
571.fRien
2
F
×××