Datasheet
REV. C
AD9631/AD9632
–13–
THEORY OF OPERATION
General
The AD9631 and AD9632 are wide bandwidth, voltage feedback
amplifiers. Since their open-loop frequency response follows the
conventional 6 dB/octave roll-off, their gain bandwidth product
is basically constant. Increasing their closed-loop gain results in
a corresponding decrease in small signal bandwidth. This can
be observed by noting the bandwidth specification between the
AD9631 (gain of +1) and AD9632 (gain of +2). The AD9631/
AD9632 typically maintain 65 degrees of phase margin. This
high margin minimizes the effects of signal and noise peaking.
Feedback Resistor Choice
The value of the feedback resistor is critical for optimum perfor-
mance on the AD9631 (gain of +1) and less critical as the gain
increases. Therefore, this section is specifically targeted at the
AD9631.
At minimum stable gain (+1), the AD9631 provides optimum
dynamic performance with R
F
= 140 W. This resistor acts as a
parasitic suppressor only against damped RF oscillations that
can occur due to lead (input, feedback) inductance and parasitic
capacitance. This value of R
F
provides the best combination of
wide bandwidth, low parasitic peaking, and fast settling time.
In fact, for the same reasons, a 100 W–130 W resistor should be
placed in series with the positive input for other AD9631 noninver-
ting and all AD9631 inverting configurations. The correct
connection is shown in Figures 3 and 4.
AD9631/
AD9632
+V
S
–V
S
100–130
R
TERM
R
IN
V
IN
R
G
0.1F
10F
0.1F
10F
R
F
V
OUT
G = 1 +
R
F
R
G
Figure 3. Noninverting Operation
AD9631/
AD9632
+V
S
–V
S
100–130
R
TERM
R
IN
R
G
0.1F
10F
0.1F
10F
R
F
V
OUT
V
IN
G = –
R
F
R
G
Figure 4. Inverting Operation
When the AD9631 is used in the transimpedance (I to V) mode,
such as in photodiode detection, the value of R
F
and diode capaci-
tance (C
I
) are usually known. Generally, the value of R
F
selected
will be in the kW range, and a shunt capacitor (C
F
) across R
F
will
be required to maintain good amplifier stability. The value of
C
F
required to maintain optimal flatness (<1 dB peaking) and
settling time can be estimated as
CCR R
FOIF OF
@
()
[]
21
22
1
2
–/
where
w
O
is equal to the unity gain bandwidth product of the
amplifier in rad/sec, and C
I
is the equivalent total input
capacitance at the inverting input. Typically
w
O
= 800 10
6
rad/sec (see TPC 15).
As an example, choosing R
F
= 10 kW and C
I
= 5 pF requires C
F
to be 1.1 pF (Note: C
I
includes both source and parasitic circuit
capacitance). The bandwidth of the amplifier can be estimated
using the C
F
calculated as
f
RC
d
FF
3
16
2
@
.
AD9631
R
F
V
OUT
C
I
I
I
C
F
Figure 5. Transimpedance Configuration
For general voltage gain applications, the amplifier bandwidth
can be closely estimated as
f
21R/R
3dB
FG
@
+
()
O
This estimation loses accuracy for gains of +2/–1 or lower due
to the amplifier’s damping factor. For these “low gain” cases,
the bandwidth will actually extend beyond the calculated value
(see TPCs 13 and 25).
As a general rule, capacitor C
F
will not be required if
RR C
NG
FG I
()
¥£
4
O
where NG is the noise gain (1 + R
F
/R
G
) of the circuit. For most
voltage gain applications, this should be the case.