Datasheet

AD539
Rev. B | Page 12 of 20
APPLICATIONS INFORMATION
BASIC MULTIPLIER CONNECTIONS
Figure 20 shows the connections for the standard dual-channel
multiplier, using op amps to provide useful output power and
the AD539 feedback resistors to achieve accurate scaling. The
transfer function for each channel is
V
W
= −V
X
V
Y
where the inputs and outputs are expressed in volts (see the
Transfer Function section).
At the nominal full-scale inputs of V
X
= 3 V and V
Y
= ±2 V, the
full-scale outputs are ±6 V. Depending on the choice of op amp,
their supply voltages may need to be about 2 V more than the
peak output. Thus, supplies of at least ±8 V are required; the
AD539 can share these supplies. Higher outputs are possible if
V
X
and V
Y
are driven to their peak values of +3.2 V and ±4.2 V,
respectively, when the peak output is ±13.4 V. This requires
operating the op amps at supplies of ±15 V. Under these condi-
tions, it is advisable to reduce the supplies to the AD539 to
±7.5 V to limit its power dissipation; however, with some form
of heat-sinking, it is permissible to operate the AD539 directly
from ±15 V supplies.
09679-020
1
2
3
4
16
15
14
13
5 12
6 11
7 10
8 9
AD539
V
X
HF COMP
V
Y1
V
Y2
+V
S
+V
S
–V
S
INPUT
COMMON
OUTPUT
COMMON
NOTES
1. ALL DECOUPLING CAPACITORS ARE 0.47µF CERAMIC.
BASE
COMMON
W1
Z1
CHAN1
OUTPUT
CHAN2
OUTPUT
Z2
W2
C
C
= 3nF
V
X
V
W1
=
–V
X
V
Y1
V
W2
=
–V
X
V
Y2
NC
NC
C
F
C
F
+V
S
–V
S
–V
S
–V
S
V
Y1
V
Y2
Figure 20. Standard Dual-Channel Multiplier
(16-Lead SBDIP and PDIP Shown)
Viewed as a voltage-controlled amplifier, the decibel gain is simply
G = 20 log V
X
where V
X
is expressed in volts. This results in a gain of 10 dB at
V
X
= 3.162 V, 0 dB at V
X
= 1 V, −20 dB at V
X
= 0.1 V, and so on.
In many ac applications, the output offset voltage (for V
X
= 0 V
or V
Y
= 0 V) is not a major concern; however, it can be elimi-
nated using the offset nulling method recommended for the
particular op amp, with V
X
= V
Y
= 0 V.
At small values of V
X
, the offset voltage of the control channel
degrades the gain/loss accuracy. For example, a ±1 mV offset
uncertainty causes the nominal 40 dB attenuation at V
X
=
0.01 V to range from 39.2 dB to 40.9 dB. Figure 4 shows the
maximum gain error boundaries based on the guaranteed
control channel offset voltages of ±2 mV for the AD539K and
±4 mV for the AD539J. These curves include all scaling errors
and apply to all configurations using the internal feedback
resistors (W1 and W2 or, alternatively, Z1 and Z2).
Distortion is a function of the signal input level (V
Y
) and the
control input (V
X
). It is also a function of frequency, although
in practice, the op amp generates most of the distortion at frequen-
cies above 100 kHz. Figure 5 shows typical results at f = 10 kHz
as a function of V
X
with V
Y
= 0.5 V rms and 1.5 V rms.
In some cases, it may be desirable to alter the scaling. This can
be achieved in several ways. One option is to use both the Z and
W feedback resistors (see Figure 18) in parallel, in which case
V
W
= −V
X
V
Y
/2. This may be preferable where the output swing
must be held at ±3 V FS (±6.75 peak), for example, to allow the use
of reduced supply voltages for the op amps. Alternatively, the
gain can be doubled by connecting both channels in parallel and
using only a single feedback resistor, in which case V
W
= −2V
X
Y
Y
and the full-scale output is ±12 V. Another option is to insert a
resistor in series with the control channel input, permitting the
use of a large (for example, 0 V to 10 V) control voltage. A
disadvantage of this scheme is the need to adjust this resistor to
accommodate the tolerance of the nominal 500 Ω input resistance
at Pin 1, V
X
. The signal channel inputs can also be resistively
attenuated to permit operation at higher values of V
Y
, in which
case it may often be possible to partially compensate for the
response roll-off of the op amp by adding a capacitor across the
upper arm of this attenuator.
Signal Channel AC and Transient Response
The HF response is dependent almost entirely on the op amp.
Note that the noise gain for the op amp in Figure 20 is determined
by the value of the feedback resistor (6 kΩ) and the 1.25 kΩ
control-bias resistors (see Figure 18). Op amps with provision
for external frequency compensation should be compensated
for a closed-loop gain of 6.
The layout of the circuit components is very important if low
feedthrough and flat response at low values of V
X
is to be
maintained (see the General Recommendations section).
For wide bandwidth applications requiring an output voltage
swing greater than ±1 V, the LH0032 hybrid op amp is recom-
mended. Figure 6 shows the HF response of the circuit of Figure 20
usin
g this amplifier with V
Y
= 1 V rms and other conditions
as shown in Table 4. C
F
was adjusted for 1 dB peaking at V
X
= 1
V; the −3 dB bandwidth exceeds 25 MHz. The effect of signal
feedthrough on the response becomes apparent at V
X
= 0.01 V.
The minimum feedthrough results when V
X
is taken slightly
negative to ensure that the residual control channel offset is
exceeded and the dc gain is reliably zero. Measurements show
that the feedthrough can be held to −90 dB relative to full
output at low frequencies and to −60 dB up to 20 MHz with
careful board layout. The corresponding pulse response is
shown in Figure 7 for a signal input of V
Y
of ±1 V and two
values of V
X
(3 V and 0.1 V).